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LTC3769MPUF#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30P31-P32
LTC3769
16
3769fa
For more information www.linear.com/LTC3769
APPLICATIONS INFORMATION
Sense Resistor Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 2a. R
SENSE
is chosen based on the required
output current.
The current comparator has a maximum threshold
V
SENSE(MAX)
. When the ILIM pin is grounded, floating or
tied to INTV
CC
, the maximum threshold is set to 50mV,
75mV or 100mV, respectively. The current comparator
threshold sets the peak of the inductor current, yielding
a maximum average inductor current, I
MAX
, equal to the
peak value less half the peak-to-peak ripple current, ΔI
L
.
To calculate the sense resistor value, use the equation:
R
SENSE
=
V
SENSE(MAX)
I
MAX
+
ΔI
L
2
The actual value of I
MAX
depends on the required output
current I
OUT(MAX)
and can be calculated using:
I
MAX
=
I
OUT(MAX)
•
V
OUT
V
IN
When using the controller in low V
IN
and very high voltage
output applications, the maximum inductor current and
correspondingly the maximum output current level will
be reduced due to the internal compensation required to
meet stability criterion for boost regulators operating at
greater than 50% duty factor. A curve is provided in the
Typical Performance Characteristics section to estimate
this reduction in peak inductor current level depending
upon the operating duty factor.
Inductor DCR Sensing
For applications requiring the highest possible efficiency
at high load currents, the LTC3769 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 2b. The DCR of the inductor can be less than 1mΩ
for high current inductors. In a high current application
requiring such an inductor, conduction loss through a
sense resistor could reduce the efficiency by a few percent
compared to DCR sensing.
If the external R1||R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature. Consult
the manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the induct-
or value calculation section, the target sense resistor
value is:
R
SENSE(EQUIV)
=
V
SENSE(MAX)
I
MAX
+
ΔI
L
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the maximum current sense threshold
(V
SENSE(MAX)
).
Next, determine the DCR of the inductor. Where provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of resistance, which is approximately 0.4%/°C.
A conservative value for the maximum inductor temperature
(T
L(MAX)
) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor value, use the divider ratio:
R
D
=
R
SENSE(EQUIV)
DCR
MAX
at T
L(MAX)
C1 is usually selected to be in the range of 0.1μF to 0.47μF.
This forces R1|| R2 to around 2k, reducing error that might
have been caused by the SENSE
–
pin’s ±1μA current.
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
R1||R2=
L
(DCR at 20°C)•C1
LTC3769
17
3769fa
For more information www.linear.com/LTC3769
APPLICATIONS INFORMATION
The sense resistor values are:
R1=
R1||R2
R
D
; R2 =
R1•R
D
1− R
D
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at V
IN
= 1/2V
OUT
:
P
LOSS_R1
=
(V
OUT
− V
IN
)• V
IN
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduc
-
tion losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
Inductor Value Calculation
The
operating frequency and inductor selection are in
-
terrelated in that higher operating frequencies allow the
use of smaller inductor and capacitor values. Why would
anyone ever
choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge and switching losses. Also, at
higher frequency the duty cycle of body diode conduction
is higher, which results in lower efficiency. In addition to
this basic trade-off, the effect of inductor value on ripple
current and low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current ΔI
L
decreases with higher
inductance or frequency and increases with higher V
IN
:
ΔI
L
=
V
IN
f •L
1−
V
IN
V
OUT
⎛
⎝
⎜
⎞
⎠
⎟
Accepting larger values of ΔI
L
allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ΔI
L
= 0.3(I
MAX
). The maximum
ΔI
L
occurs at V
IN
= 1/2V
OUT
.
The inductor value also has secondary effects. The tran-
sition to Burst Mode operation begins when the average
inductor
current required results in a
peak current below
25% of the current limit determined by R
SENSE
. Lower
inductor values (higher ΔI
L
) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease. Once the value of L is known, an
inductor with low DCR and low core losses should be
selected.
Power MOSFET Selection
Two external power MOSFETs must be selected for the
LTC3769: one N-channel MOSFET for the bottom (main)
switch, and one N-channel MOSFET for the top (synchro
-
nous) switch.
The peak-to-peak gate drive levels are set by the INT
V
CC
voltage. This voltage is typically 5.4V during start-up
(see EXTV
CC
pin connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
Pay close attention to the BV
DSS
specification for the
MOSFETs as well; many of the logic level MOSFETs are
limited to 30V or less.
Selection criteria for the power MOSFETs include the
on-resistance R
DS(ON)
, Miller capacitance C
MILLER
, input
voltage and maximum output current. Miller capacitance,
C
MILLER
, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturer’s data
sheet. C
MILLER
is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result
is then multiplied by the ratio of the application applied
VDS to the gate charge curve specified VDS. When the IC
is operating in continuous mode, the duty cycles for the
top and bottom MOSFETs are given by:
Main Switch Duty Cycle=
V
OUT
− V
IN
V
OUT
Synchronous SwitchDuty Cycle =
V
IN
V
OUT
LTC3769
18
3769fa
For more information www.linear.com/LTC3769
APPLICATIONS INFORMATION
If the maximum output current is I
OUT(MAX)
the MOSFET
power dissipation at maximum output current is given by:
P
MAIN
=
(V
OUT
V
IN
)V
OUT
V
2
IN
•
I
OUT(MAX)
2
• 1+
( )
• R
DS(ON)
+k • V
OUT
3
•
I
OUT(MAX)
V
IN
• •C
MILLER
•
f
P
SYNC
=
V
IN
V
OUT
I
OUT(MAX)
2
• 1+
( )
•R
DS(ON)
where d is the temperature dependency of R
DS(ON)
. The
constant k, which accounts for the loss caused by reverse
recovery current, is inversely proportional to the gate drive
current and has an empirical value of 1.7.
Both MOSFETs have I
2
R losses while the bottom N-channel
equation includes an additional term for transition losses,
which are highest at low input voltages. For high V
IN
the
high current efficiency generally improves with larger
MOSFETs, while for low V
IN
the transition losses rapidly
increase to the point that the use of a higher R
DS(ON)
device
with lower C
MILLER
actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the bottom switch duty factor is low or dur
-
ing overvoltage when the synchronous switch is on close
to 100% of the period.
The term (1+ d) is generally given for a MOSFET in the
form of a normalized R
DS(ON)
vs Temperature curve, but
d = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
C
IN
and C
OUT
Selection
The input ripple current in a boost converter is relatively
low (compared with the output ripple current), because this
current is continuous. The input capacitor C
IN
voltage rating
should comfortably exceed the maximum input voltage.
Although ceramic capacitors can be relatively tolerant of
overvoltage conditions, aluminum electrolytic capacitors
are not. Be sure to characterize the input voltage for any
possible overvoltage transients that could apply excess
stress to the input capacitors.
The value of C
IN
is a function of the source impedance, and
in general, the higher the source impedance, the higher the
required input capacitance. The required amount of input
capacitance is also greatly affected by the duty cycle. High
output current applications that also experience high duty
cycles can place great demands on the input supply, both
in terms of DC current and ripple current.
In a boost converter, the output has a discontinuous current,
so C
OUT
must be capable of reducing the output voltage
ripple. The effects of ESR (equivalent series resistance) and
the bulk capacitance must be considered when choosing
the right capacitor for a given output ripple voltage. The
steady ripple voltage due to charging and discharging
the bulk capacitance in a single phase boost converter
is given by:
V
RIPPLE
=
I
OUT(MAX )
•(V
OUT
− V
IN(MIN)
)
C
OUT
• V
OUT
• f
V
where C
OUT
is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
ΔV
ESR
= I
L(MAX)
• ESR
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings (e.g., OS-CON and POSCAP).
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LTC3769MPUF#TRPBF

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Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 60V L IQ Sync Boost Cntr
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