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AD640BPZ

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20
AD640
REV. D –9–
(see Figure 20). For the AD640, V
X
is calibrated to exactly
1 mV. The slope of the line is directly proportional to V
Y
. Base
10 logarithms are used in this context to simplify the relation-
ship to decibel values. For V
IN
= 10 V
X
, the logarithm has a
value of 1, so the output voltage is V
Y
. At V
IN
= 100 V
X
, the
output is 2 V
Y
, and so on. V
Y
can therefore be viewed either as
the Slope Voltage or as the Volts per Decade Factor.
0
V
Y
2V
Y
V
IN
= V
X
V
IN
= 10V
X
V
IN
= 100V
X
SLOPE = V
Y
ACTUAL
IDEAL
INPUT ON
LOG SCALE
ACTUAL
IDEAL
V
Y
LOG (V
IN
/V
X
)
Figure 20. Basic DC Transfer Function of the AD640
The AD640 conforms to Equation (1) except that its two out-
puts are in the form of currents, rather than voltages:
I
OUT
= I
Y
LOG (V
IN
/V
X
) Equation (2)
I
Y
the Slope Current, is 1 mA. The current output can readily be
converted to a voltage with a slope of 1 V/decade, for example,
using one of the 1 kΩ resistors provided for this purpose, in
conjunction with an op amp, as shown in Figure 21.
1115 14 13 12
6
7
8
9
10
SIG
+OUT
LOG
COM
LOG
OUT
+V
S
–V
S
ITC BL2
SIG
–OUT
AD640
C1
330pF
AD844
R1
48.7V
R2
1mA PER
DECADE
OUTPUT VOLTAGE
1V PER DECADE
FOR R2 = 1kV
100mV PER dB
for R2 = 2kV
Figure 21. Using an External Op Amp to Convert the
AD640 Output Current to a Buffered Voltage Output
Intercept Stabilization
Internally, the intercept voltage is a fraction of the thermal volt-
age kT/q, that is, V
X
= V
XO
T/T
O
, where V
XO
is the value of V
X
at a reference temperature T
O
. So the uncorrected transfer
function has the form
I
OUT
= I
Y
LOG (V
IN
T
O
/V
XO
T) Equation (3)
Now, if the amplitude of the signal input V
IN
could somehow be
rendered PTAT, the intercept would be stable with tempera-
ture, since the temperature dependence in both the numerator
and denominator of the logarithmic argument would cancel.
This is what is actually achieved by interposing the on-chip
attenuator, which has the necessary temperature dependence to
cause the input to the first stage to vary in proportion to abso-
lute temperature. The end limits of the dynamic range are now
totally independent of temperature. Consequently, this is the
preferred method of intercept stabilization for applications
where the input signal is sufficiently large.
When the attenuator is not used, the PTAT variation in V
X
will result in the intercept being temperature dependent. Near
300K (27°C) it will vary by 20 LOG (301/300) dB/°C, about
0.03 dB/°C. Unless corrected, the whole output function would
drift up or down by this amount with changes in temperature. In
the AD640 a temperature compensating current I
Y
LOG(T/T
O
)
is added to the output. This effectively maintains a constant
intercept V
XO
. This correction is active in the default state (Pin
8 open circuited). When using the attenuator, Pin 8 should be
grounded, which disables the compensation current. The drift
term needs to be compensated only once; when the outputs of
two AD540s are summed, Pin 8 should be grounded on at least
one of the two devices (both if the attenuator is used).
Conversion Range
Practical logarithmic converters have an upper and lower limit
on the input, beyond which errors increase rapidly. The upper
limit occurs when the first stage in the chain is driven into limit-
ing. Above this, no further increase in the output can occur and
the transfer function flattens off. The lower limit arises because
a finite number of stages provide finite gain, and therefore at
low signal levels the system becomes a simple linear amplifier.
Note that this lower limit is not determined by the intercept
voltage, V
X
; it can occur either above or below V
X
, depending
on the design. When using two AD640s in cascade, input offset
voltage and wideband noise are the major limitations to low
level accuracy. Offset can be eliminated in various ways. Noise
can only be reduced by lowering the system bandwidth, using a
filter between the two devices.
EFFECT OF WAVEFORM ON INTERCEPT
The absolute value response of the AD640 allows inputs of
either polarity to be accepted. Thus, the logarithmic output in
response to an amplitude-symmetric square wave is a steady
value. For a sinusoidal input the fluctuating output current will
usually be low-pass filtered to extract the baseband signal. The
unfiltered output is at twice the carrier frequency, simplifying the
design of this filter when the video bandwidth must be maxi-
mized. The averaged output depends on waveform in a roughly
analogous way to waveform dependence of rms value. The effect
is to change the apparent intercept voltage. The intercept volt-
age appears to be doubled for a sinusoidal input, that is, the
averaged output in response to a sine wave of amplitude (not rms
value) of 20 mV would be the same as for a dc or square wave
input of 10 mV. Other waveforms will result in different inter-
cept factors. An amplitude-symmetric-rectangular waveform
has the same intercept as a dc input, while the average of a
baseband unipolar pulse can be determined by multiplying the
response to a dc input of the same amplitude by the duty cycle.
It is important to understand that in responding to pulsed RF
signals it is the waveform of the carrier (usually sinusoidal) not
the modulation envelope, that determines the effective intercept
voltage. Table I shows the effective intercept and resulting deci-
bel offset for commonly occurring waveforms. The input wave-
form does not affect the slope of the transfer function. Figure 22
shows the absolute deviation from the ideal response of cascaded
AD640s for three common waveforms at input levels from
–80 dBV to –10 dBV. The measured sine wave and triwave
responses are 6 dB and 8.7 dB, respectively, below the square
wave response—in agreement with theory.
AD640
REV. D–10–
Table I.
Input Peak Intercept Error (Relative
Waveform or RMS Factor to a DC Input)
Square Wave Either 1 0.00 dB
Sine Wave Peak 2 –6.02 dB
Sine Wave rms 1.414(√2) –3.01 dB
Triwave Peak 2.718 (e) –8.68 dB
Triwave rms 1.569(e/√3) –3.91 dB
Gaussian Noise rms 1.887 –5.52 dB
Logarithmic Conformance and Waveform
The waveform also affects the ripple, or periodic deviation from
an ideal logarithmic response. The ripple is greatest for dc or
square wave inputs because every value of the input voltage
maps to a single location on the transfer function and thus
traces out the full nonlinearities in the logarithmic response.
By contrast, a general time varying signal has a continuum of
values within each cycle of its waveform. The averaged output is
thereby “smoothed” because the periodic deviations away from
the ideal response, as the waveform “sweeps over” the transfer
function, tend to cancel. This smoothing effect is greatest for a
triwave input, as demonstrated in Figure 22.
INPUT AMPLITUDE IN dB ABOVE 1V, AT 10kHz
2
–10
–80
DEVIATION FROM EXACT LOGARITHMIC
TRANSFER FUNCTION – dB
–8
–6
–4
–2
0
–70 –60 –50 –40 –30 –20 –10
SQUARE WAVE INPUT
SINE WAVE INPUT
TRIWAVE INPUT
Figure 22. Deviation from Exact Logarithmic Transfer
Function for Two Cascaded AD640s, Showing Effect of
Waveform on Calibration and Linearity
INPUT AMPLITUDE IN dB ABOVE 1V, AT 10kHz
2
–10
DEVIATION FROM EXACT LOGARITHMIC
TRANSFER FUNCTION – dB
–8
–6
–4
–2
0
–70 –60 –50 –40 –30 –20 –10
SQUARE WAVE INPUT
SINE WAVE INPUT
TRIWAVE INPUT
–12
4
Figure 23. Deviation from Exact Logarithmic Transfer
Function for a Single AD640; Compare Low Level
Response with that of Figure 22
The accuracy at low signal inputs is also waveform dependent.
The detectors are not perfect absolute value circuits, having a
sharp “corner” near zero; in fact they become parabolic at low
levels and behave as if there were a dead zone. Consequently,
the output tends to be higher than ideal. When there are enough
stages in the system, as when two AD640s are connected in
cascade, most detectors will be adequately loaded due to the
high overall gain, but a single AD640 does not have sufficient
gain to maintain high accuracy for low level sine wave or triwave
inputs. Figure 23 shows the absolute deviation from calibration
for the same three waveforms for a single AD640. For inputs
between –10 dBV and –40 dBV the vertical displacement of the
traces for the various waveforms remains in agreement with the
predicted dependence, but significant calibration errors arise at
low signal levels.
SIGNAL MAGNITUDE
AD640 is a calibrated device. It is, therefore, important to be
clear in specifying the signal magnitude under all waveform
conditions. For dc or square wave inputs there is, of course, no
ambiguity. Bounded periodic signals, such as sinusoids and
triwaves, can be specified in terms of their simple amplitude
(peak value) or alternatively by their rms value (which is a mea-
sure of power when the impedance is specified). It is generally bet-
ter to define this type of signal in terms of its amplitude because
the AD640 response is a consequence of the input voltage, not
power. However, provided that the appropriate value of inter-
cept for a specific waveform is observed, rms measures may be
used. Random waveforms can only be specified in terms of rms
value because their peak value may be unbounded, as is the case
for Gaussian noise. These must be treated on a case-by-case
basis. The effective intercept given in Table I should be used for
Gaussian noise inputs.
On the other hand, for bounded signals the amplitude can be
expressed either in volts or dBV (decibels relative to 1 V). For
example, a sine wave or triwave of 1 mV amplitude can also be
defined as an input of –60 dBV, one of 100 mV amplitude as
–20 dBV, and so on. RMS value is usually expressed in dBm
(decibels above 1 mW) for a specified impedance level. Through-
out this data sheet we assume a 50
Ω
environment, the customary
impedance level for high speed systems, when referring to signal power
in dBm. Bearing in mind the above discussion of the effect of
waveform on the intercept calibration of the AD640, it will be
apparent that a sine wave at a power of, say, –10 dBm will not
produce the same output as a triwave or square wave of the
same power. Thus, a sine wave at a power level of –10 dBm has
an rms value of 70.7 mV or an amplitude of 100 mV (that is, √2
times as large, the ratio of amplitude to rms value for a sine
wave), while a triwave of the same power has an amplitude
which is √3 or 1.73 times its rms value, or 122.5 mV.
“Intercept” and “Logarithmic Offset”
If the signals are expressed in dBV, we can write the output in a
simpler form, as
I
OUT
= 50
µ
A (Input
dBV
– X
dBV
) Equation (4)
where Input
dBV
is the input voltage amplitude (not rms) in dBV
and X
dBV
is the appropriate value of the intercept (for a given
waveform) in dBV. This form shows more clearly why the intercept
is often referred to as the logarithmic offset. For dc or square
wave inputs, V
X
is 1 mV so the numerical value of X
dBV
is –60,
and Equation (4) becomes
AD640
REV. D –11–
I
OUT
= 50
µ
A (Input
dBV
+ 60) Equation (5)
Alternatively, for a sinusoidal input measured in dBm (power in
dB above 1 mW in a 50 Ω system) the output can be written
I
OUT
= 50
µ
A (Input
dBm
+ 44) Equation (6)
because the intercept for a sine wave expressed in volts rms is at
1.414 mV (from Table I) or –44 dBm.
OPERATION OF A SINGLE AD640
Figure 24 shows the basic connections for a single device, using
100 Ω load resistors. Output A is a negative going voltage with a
slope of –100 mV per decade; output B is positive going with a
slope of +100 mV per decade. For applications where absolute
calibration of the intercept is essential, the main output (from
LOG OUT, Pin 14) should be used; the LOG COM output can
then be grounded. To evaluate the demodulation response, a
simple low-pass output filter having a time constant of roughly
500 µs (3 dB corner of 320 Hz) is provided by a 4.7 µF (–20%
+80%) ceramic capacitor (Erie type RPE117-Z5U-475-K50V)
placed across the load. A DVM may be used to measure the
averaged output in verification tests. The voltage compliance at
Pins 13 and 14 extends from 0.3 V below ground up to 1 V
below +V
S
. Since the current into Pin 14 is from –0.2 mA at
zero signal to +2.3 mA when fully limited (dc input of >300 mV)
the output never drops below –230 mV. On the other hand, the
current out of Pin 13 ranges from 0.2 mA to +2.3 mA, and if
desired, a load resistor of up to 2 kΩ can be used on this output;
the slope would then be 2 V per decade. Use of the LOG COM
output in this way provides a numerically correct decibel read-
ing on a DVM (+100 mV = +1.00 dB).
Board layout is very important. The AD640 has both high gain
and wide bandwidth; therefore every signal path must be very
carefully considered. A high quality ground plane is essential,
but it should not be assumed that it behaves as an equipotential
plane. Even though the application may only call for modest
bandwidth, each of the three differential signal interface pairs
(SIG IN, Pins 1 and 20, SIG OUT, Pins 10 and 11, and LOG,
Pins 13 and 14) must have their own “starred” ground points to
avoid oscillation at low signal levels (where the gain is highest).
Unused pins (excluding Pins 8, 10 and 11) such as the attenua-
tor and applications resistors should be grounded close to the
package edge. BL1 (Pin 6) and BL2 (Pin 9) are internal bias
lines a volt or two above the –V
S
node; access is provided solely
for the addition of decoupling capacitors, which should be con-
nected exactly as shown (not all of them connect to the ground).
Use low impedance ceramic 0.1 µF capacitors (for example,
Erie RPE113-Z5U-105-K50V). Ferrite beads may be used
instead of supply decoupling resistors in cases where the supply
voltage is low.
Active Current-to-Voltage Conversion
The compliance at LOG OUT limits the available output volt-
age swing. The output of the AD640 may be converted to a
larger, buffered output voltage by the addition of an operational
amplifier connected as a current-to-voltage (transresistance)
stage, as shown in Figure 21. Using a 2 kΩ feedback resistor
(R2) the 50 µA/dB output at LOG OUT is converted to a volt-
age having a slope of +100 mV/dB, that is, 2 V per decade. This
output ranges from roughly –0.4 V for zero signal inputs to the
AD640, crosses zero at a dc input of precisely +1 mV (or
–1 mV) and is +4 V for a dc input of 100 mV. A passive
prefilter, formed by R1 and C1, minimizes the high frequency
energy conveyed to the op amp. The corner frequency is here
shown as 10 MHz. The AD844 is recommended for this appli-
cation because of its excellent performance in transresistance
modes. Its bandwidth of 35 MHz (with the 2 kΩ feedback resis-
tor) will exceed the baseband response of the system in most
applications. For lower bandwidth applications other op amps
and multipole active filters may be substituted (see, for example,
Figure 32 in the APPLICATIONS section).
Effect of Frequency on Calibration
The slope and intercept of the AD640 are calibrated during
manufacture using a 2 kHz square wave input. Calibration de-
pends on the gain of each stage being 10 dB. When the input
frequency is an appreciable fraction of the 350 MHz bandwidth
of the amplifier stages, their gain becomes imprecise and the
logarithmic slope and intercept are no longer fully calibrated.
However, the AD640 can provide very stable operation at fre-
quencies up to about one half the 3 dB frequency of the ampli-
fier stages. Figure 10 shows the averaged output current versus
input level at 30 MHz, 60 MHz, 90 MHz and 120 MHz. Fig-
ure 11 shows the absolute error in the response at 60 MHz and
at temperatures of –55°C, +25°C and +125°C. Figure 12 shows
the variation in the slope current, and Figure 13 shows the
variation in the intercept level (sinusoidal input) versus frequency.
If absolute calibration is essential, or some other value of slope
or intercept is required, there will usually be some point in the
user’s system at which an adjustment may be easily introduced.
For example, the 5% slope deficit at 30 MHz (see Figure 12)
may be restored by a 5% increase in the value of the load resis-
tor in the passive loading scheme shown in Figure 24, or by
inserting a trim potentiometer of 100 Ω in series with the feed-
back resistor in the scheme shown in Figure 21. The intercept
NC
R
LA
100V
0.1%
4.7mF
R
LB
100V
0.1%
4.7mF
OUTPUT A
OUTPUT B
NC
15 13141619 18 17 1112
20
6
8
753
4 1091
2
SIG
+IN
ATN
OUT
CKT
COM
RG1 RG0 RG2 LOG
OUT
LOG
COM
+V
S
SIG
+OUT
SIG
–IN
ATN
LO
ATN
COM
BL1 BL2ITC
–V
S
SIG
–OUT
1kV 1kV
ATN
COM
ATN
IN
AD640
NC
4.7V
10V
+5V
–5V
OPTIONAL
TERMINATION
RESISTOR
SIGNAL
INPUT
DENOTES A SHORT, DIRECT CONNECTION
TO THE GROUND PLANE.
ALL UNMARKED CAPACITORS ARE
0.1mF CERAMIC (SEE TEXT)
OPTIONAL
OFFSET BALANCE
RESISTOR
NC = NO CONNECT
Figure 24. Connections for a Single AD640 to Verify Basic Performance
P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20

AD640BPZ

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