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LTC3717EUH-1#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P20
10
LTC3717-1
sn37171 37171fs
APPLICATIO S I FOR ATIO
WUUU
D
V
V
D
VV
V
TOP
OUT
IN
BOT
IN OUT
IN
=
=
–
The resulting power dissipation in the MOSFETs at maxi-
mum output current are:
P
TOP
= D
TOP
I
OUT(MAX)
2
ρ
T(TOP)
R
DS(ON)(MAX)
+ k V
IN
2
I
OUT(MAX)
C
RSS
f
P
BOT
= D
BOT
I
OUT(MAX)
2
ρ
T(BOT)
R
DS(ON)(MAX)
Both MOSFETs have I
2
R losses and the top MOSFET
includes an additional term for transition losses, which are
largest at high input voltages. The constant k = 1.7A
–1
can
be used to estimate the amount of transition loss. The
bottom MOSFET losses are greatest when the bottom duty
cycle is near 100%, during a short-circuit or at high input
voltage.
Operating Frequency
The choice of operating frequency is a tradeoff between
efficiency and component size. Low frequency operation
improves efficiency by reducing MOSFET switching losses
but requires larger inductance and/or capacitance in order
to maintain low output ripple voltage.
The operating frequency of LTC3717-1 applications is
determined implicitly by the one-shot timer that controls
the on-time t
ON
of the top MOSFET switch. The on-time is
set by the current into the I
ON
pin and the voltage at the V
ON
pin according to:
t
V
I
pF
ON
VON
ION
= ()10
Tying a resistor R
ON
from V
IN
to the I
ON
pin yields an on-
time inversely proportional to V
IN
. For a step-down
converter, this results in approximately constant fre-
quency operation as the input supply varies:
f
V
VR pF
Hz
OUT
VON ON
=
[]
()10
To hold frequency constant during output voltage changes,
tie the V
ON
pin to V
OUT
. The V
ON
pin has internal clamps
that limit its input to the one-shot timer. If the pin is tied
below 0.7V, the input to the one-shot is clamped at 0.7V.
Similarly, if the pin is tied above 2.4V, the input is clamped
at 2.4V.
Because the voltage at the I
ON
pin is about 0.7V, the
current into this pin is not exactly inversely proportional to
V
IN
, especially in applications with lower input voltages.
To account for the 0.7V drop on the I
ON
pin, the following
equation can be used to calculate frequency:
f
VVV
VVRpF
IN OUT
VON IN ON
=
−
()
07
10
.•
•• ( )
To correct for this error, an additional resistor R
ON2
connected from the I
ON
pin to the 5V INTV
CC
supply will
further stabilize the frequency.
R
V
V
R
ON ON2
5
07
=
.
Changes in the load current magnitude will also cause
frequency shift. Parasitic resistance in the MOSFET
switches and inductor reduce the effective voltage across
the inductance, resulting in increased duty cycle as the
load current increases. By lengthening the on-time slightly
as current increases, constant frequency operation can be
maintained. This is accomplished with a resistive divider
from the I
TH
pin to the V
ON
pin and V
OUT
. The values
required will depend on the parasitic resistances in the
specific application. A good starting point is to feed about
Figure 2. R
DS(ON)
vs. Temperature
JUNCTION TEMPERATURE (°C)
–50
ρ
T
NORMALIZED ON-RESISTANCE
1.0
1.5
150
37171 F02
0.5
0
0
50
100
2.0
11
LTC3717-1
sn37171 37171fs
25% of the voltage change at the I
TH
pin to the V
ON
pin as
shown in Figure 3a. Place capacitance on the V
ON
pin to
filter out the I
TH
variations at the switching frequency. The
resistor load on I
TH
reduces the DC gain of the error amp
and degrades load regulation, which can be avoided by
using the PNP emitter follower of Figure 3b.
Inductor L1 Selection
Given the desired input and output voltages, the inductor
value and operating frequency determine the ripple
current:
∆=
−
I
V
fL
V
V
L
OUT OUT
IN
1
Lower ripple current reduces cores losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Highest efficiency operation is obtained at low
frequency with small ripple current. However, achieving
this requires a large inductor. There is a tradeoff between
component size, efficiency and operating frequency.
A reasonable starting point is to choose a ripple current
that is about 40% of I
OUT(MAX)
. The largest ripple current
occurs at the highest V
IN
. To guarantee that ripple current
does not exceed a specified maximum, the inductance
should be chosen according to:
L
V
fI
V
V
OUT
L MAX
OUT
IN MAX
=
∆
−
() ()
1
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ
®
cores. A variety of inductors designed for high
current, low voltage applications are available from manu-
facturers such as Sumida, Panasonic, Coiltronics, Coil-
craft and Toko.
Schottky Diode D1, D2 Selection
The Schottky diodes, D1 and D2, shown in Figure 1
conduct during the dead time between the conduction of
the power MOSFET switches. It is intended to prevent the
body diodes of the top and bottom MOSFETs from turning
on and storing charge during the dead time, which can
cause a modest (about 1%) efficiency loss. The diodes can
be rated for about one half to one fifth of the full load current
since they are on for only a fraction of the duty cycle. In
order for the diode to be effective, the inductance between
it and the bottom MOSFET must be as small as possible,
mandating that these components be placed adjacently.
The diodes can be omitted if the efficiency loss is tolerable.
C
IN
and C
OUT
Selection
The input capacitance C
IN
is required to filter the square
wave current at the drain of the top MOSFET. Use a low
ESR capacitor sized to handle the maximum RMS current.
II
V
V
V
V
RMS OUT MAX
OUT
IN
IN
OUT
≅
()
–1
This formula has a maximum at V
IN
= 2V
OUT
, where
I
RMS
= I
OUT(MAX)
/2. This simple worst-case condition is
commonly used for design because even significant
deviations do not offer much relief. Note that ripple
current ratings from capacitor manufacturers are often
APPLICATIO S I FOR ATIO
WUUU
Kool Mµ is a registered trademark of Magnetics, Inc.
C
VON
0.01µF
R
VON2
100k
R
VON1
30k
C
C
V
OUT
R
C
V
ON
I
TH
LTC3717-1
C
VON
0.01µF
R
VON2
10k
Q1
2N5087
R
VON1
3k
10k
C
C
37171 F03
V
OUT
INTV
CC
R
C
V
ON
I
TH
LTC3717-1
(3a) (3b)
Figure 3. Adjusting Frequency Shift with Load Current Changes
12
LTC3717-1
sn37171 37171fs
based on only 2000 hours of life which makes it advisable
to derate the capacitor.
The selection of C
OUT
is primarily determined by the ESR
required to minimize voltage ripple and load step
transients. The output ripple ∆V
OUT
is approximately
bounded by:
∆≤∆ +
V I ESR
fC
OUT L
OUT
1
8
Since ∆I
L
increases with input voltage, the output ripple is
highest at maximum input voltage. Typically, once the ESR
requirement is satisfied, the capacitance is adequate for
filtering and has the necessary RMS current rating.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Special polymer capacitors offer very low ESR
but have lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density
but it is important to only use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR, but
can be used in cost-sensitive applications providing that
consideration is given to ripple current ratings and long
term reliability. Ceramic capacitors have excellent low
ESR characteristics but can have a high voltage coeffi-
cient and audible piezoelectric effects. The high Q of
ceramic capacitors with trace inductance can also lead to
signifi
cant ringing. When used as input capacitors, care
must be taken to ensure that ringing from inrush currents
and switching does not pose an overvoltage hazard to the
power switches and controller. To dampen input voltage
transients, add a small 5µF to 50µF aluminum electrolytic
capacitor with an ESR in the range of 0.5Ω to 2Ω. High
performance through-hole capacitors may also be used,
but an additional ceramic capacitor in parallel is recom-
mended to reduce the effect of their lead inductance.
Top MOSFET Driver Supply (C
B
, D
B
)
An external bootstrap capacitor C
B
connected to the BOOST
pin supplies the gate drive voltage for the topside MOSFET.
This capacitor is charged through diode D
B
from INTV
CC
when the switch node is low. When the top MOSFET turns
APPLICATIO S I FOR ATIO
WUUU
on, the switch node rises to V
IN
and the BOOST pin rises
to approximately V
IN
+ INTV
CC
. The boost capacitor needs
to store about 100 times the gate charge required by the
top MOSFET. In most applications a 0.1µF to 0.47µF X5R
or X7R dielectric capacitor is adequate.
Fault Condition: Current Limit
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage. In
the LTC3717-1, the maximum sense voltage is controlled
by the voltage on the V
RNG
pin. With valley current control,
the maximum sense voltage and the sense resistance
determine the maximum allowed inductor valley current.
The corresponding output current limit is:
I
V
R
I
I
V
R
I
LIMITPOSITIVE
SNS MAX
DS ON T
L
LIMITNEGATIVE
SNS MIN
DS ON T
L
=+∆
=−∆
()
()
()
()
ρ
ρ
1
2
1
2
The current limit value should be checked to ensure that
I
LIMIT(MIN)
> I
OUT(MAX)
. The minimum value of current limit
generally occurs with the largest V
IN
at the highest ambi-
ent temperature, conditions that cause the largest power
loss in the converter. Note that it is important to check for
self-consistency between the assumed MOSFET junction
temperature and the resulting value of I
LIMIT
which heats
the MOSFET switches.
Caution should be used when setting the current limit
based upon the R
DS(ON)
of the MOSFETs. The maximum
current limit is determined by the minimum MOSFET on-
resistance. Data sheets typically specify nominal and
maximum values for R
DS(ON)
, but not a minimum. A
reasonable assumption is that the minimum R
DS(ON)
lies
the same amount below the typical value as the maximum
lies above it. Consult the MOSFET manufacturer for further
guidelines.
Minimum Off-time and Dropout Operation
The minimum off-time t
OFF(MIN)
is the smallest amount of
time that the LTC3717-1 is capable of turning on the
bottom MOSFET, tripping the current comparator and
turning the MOSFET back off. This time is generally about
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LTC3717EUH-1#TRPBF

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Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators Power Supply for DDR in QFN
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