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LTC1703IG#TRPBF

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LTC1703
1703fa
drops, the FCB pin will trip and the LTC1703 will resume
continuous operation regardless of the load on the main
output. The FCB pin removes the requirement that power
must be drawn from the inductor primary in order to
extract power from the auxiliary windings. With the loop in
continuous mode, the auxiliary outputs may be loaded
without regard to the primary load. Note that if the LTC1703
is already running in continuous mode and the auxiliary
output drops due to excessive loading, no additional
action can be taken by the LTC1703 to regulate the
auxiliary output.
Accuracy Trade-Offs
The V
DS
sensing scheme used in the LTC1703 is not
particularly accurate, primarily due to uncertainty in the
R
DS(ON)
from MOSFET to MOSFET. A second error term
arises from the ringing present at the SW pin, which
causes the V
DS
to look larger than (I
LOAD
)(R
DS(ON)
) at the
beginning of QB’s on-time. These inaccuracies do not
prevent the LTC1703 current limit circuit from protecting
itself and the load from damaging overcurrent conditions,
but they do prevent the user from setting the current limit
to a tight tolerance if more than one copy of the circuit is
being built. The 50% factor in the current setting equation
above reflects the margin necessary to ensure that the
circuit will stay out of current limit at the maximum normal
load, even with a hot MOSFET that is running quite a bit
higher than its R
DS(ON)
spec.
FCB OPERATION/SECONDARY WINDINGS
The FCB pin can be used in conjunction with a secondary
winding on one side of the LTC1703 to generate a third
regulated voltage output. This output can be directly
regulated at the FCB pin. In theory, a fourth output could
be added, either unregulated or with additional external
circuitry at the FCB pin.
The extra auxiliary output is taken from a second winding
on the core of the inductor on one channel, converting it
into a transformer (Figure 13). The auxiliary output voltage
is set by the main output voltage and the turns ratio of the
extra winding to the primary winding. Load regulation at
the auxiliary output will be relatively good as long as the
main output is running in continuous mode. As the load on
the main channel drops and the LTC1703 switches to
discontinuous or Burst Mode operation, the auxiliary
output will not be able to maintain regulation, especially if
the load at the auxiliary output remains heavy.
To avoid this, the auxiliary output voltage is divided down
with a conventional feedback resistor string with the
divided auxiliary output voltage fed back to the FCB pin
(Figure 13). The FCB pin threshold is trimmed to 800mV
with 20mV of hysteresis, allowing fairly precise control of
the auxiliary voltage. If the LTC1703 is in discontinuous or
Burst Mode operation and the auxiliary output voltage
Figure 13. Regulating an Auxiliary Output with the FCB Pin
+
TG
LTC1703
BG
FCB
C
OUT
R
FCB1
+
C
OUT(AUX)
V
OUT(AUX)
1703 F13
+
C
IN
QT
QB
V
OUT
V
IN
R
FCB2
FAULT FLAG
The FAULT pin is an open-drain output that indicates if one
or both of the outputs has exceeded 15% of its pro-
grammed output voltage. FAULT includes an internal
10µA pull-up to V
CC
and does not require an external pull-
up to interface to standard logic. FAULT pulls low in
normal operation, and releases when a overvoltage fault is
detected.
When an overvoltage fault occurs, an internal latch sets
and FAULT goes high, disabling the LTC1703 until the
latch is cleared by recycling the power or pulling both
RUN/SS pins low simultaneously. Alternately, the FAULT
pin can be pulled back low externally with an open-
collector/open-drain device or an N-channel MOSFET or
NPN, which will allow the LTC1703 to resume normal
operation, but will not reset the latch. If the pull-down is
later removed, the LTC1703 will latch off again unless the
latch is reset by cycling the power or RUN/SS pins.
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LTC1703
1703fa
OPTIMIZING PERFORMANCE
2-Step Conversion
The LTC1703 is ideally suited for use in 2-step conversion
systems. 2-step systems use a primary regulator to con-
vert the input power source (batteries or AC line voltage)
to an intermediate supply voltage, often 5V. The LTC1703
then converts the intermediate voltage to the low voltage,
high current supplies required by the system. Compared
to a 1-step converter that converts a high input voltage
directly to a very low output voltage, the 2-step converter
exhibits superior transient response, smaller component
size and equivalent efficiency. Thermal management and
layout complexity are also improved with a 2-step
approach.
A typical notebook computer supply might use a 4-cell
Li-Ion battery pack as an input supply with a 15V nominal
terminal voltage. The logic circuits require 5V/3A and
3.3V/5A to power system board logic, and 2.5V/0.5A,
1.5V/2A and 1.3V/10A to power the CPU. A typical 2-step
conversion system would use a step-down switcher (per-
haps an LTC1628 or two LTC1625s) to convert 15V to 5V
and another to convert 15V to 3.3V (Figure 14). One
channel of the LTC1703 would generate the 1.3V supply
using the 3.3V supply as the input and the other channel
would generate 1.5V using the 5V supply as the input. The
corresponding 1-step system would use four similar step-
down switchers, each using 15V as the input supply and
generating one of the four output voltages. Since the 2.5V
supply represents a small fraction of the total output
power, either system can generate it from the 3.3V output
using an LDO linear regulator, without the 75% linear
efficiency making much of an impact on total system
efficiency.
Figure 14. 2-Step Conversion Block Diagram
Clearly, the 5V and 3.3V sections of the two schemes are
equivalent. The 2-step system draws additional power
from the 5V and 3.3V outputs, but the regulation tech-
niques and trade-offs at these outputs are similar. The
difference lies in the way the 1.5V and 1.3V supplies are
generated. For example, the 2-step system converts 3.3V
to 1.3V with a 39% duty cycle. During the QT on-time, the
voltage across the inductor is 2V and during the QB
on-time, the voltage is 1.3V, giving roughly symmetrical
transient response to positive and negative load steps. The
2V maximum voltage across the inductor allows the use of
a small 0.47µH inductor while keeping ripple current
under 4A (40% of the 10A maximum load). By contrast,
the 1-step converter is converting 15V to 1.3V, requiring
just a 9% duty cycle. Inductor voltages are now 13.7V
when QT is on and 1.3V when QB is on, giving vastly
different di/dt values and correspondingly skewed tran-
sient response with positive and negative current steps.
The narrow 9% duty cycle usually requires a lower switch-
ing frequency, which in turn requires a higher value
inductor and larger output capacitor. Parasitic losses due
to the large voltage swing at the source of QT cost
efficiency, eliminating any advantage the 1-step conver-
sion might have had.
Note that power dissipation in the LTC1703 portion of a
2-step circuit is lower than it would be in a typical 1-step
converter, even in cases where the 1-step converter has
higher total efficiency than the 2-step system. In a typical
microprocessor core supply regulator, for example, the
regulator is usually located right next to the CPU. In a
1-step design, all of the power dissipated by the core
regulator is right there next to the hot CPU, aggravating
thermal management. In a 2-step LTC1703 design, a
significant percentage of the power lost in the core regu-
lation system happens in the 5V or 3.3V supply, which is
usually away from the CPU. The power lost to heat in the
LTC1703 section of the system is relatively low, minimiz-
ing the heat near the CPU.
2-Step Efficiency Calculation
Calculating the efficiency of a 2-step converter system
involves some subtleties. Simply multiplying the effi-
ciency of the primary 5V or 3.3V supply by the efficiency
of the 1.5V or 1.3V supply underestimates the actual
V
BAT
15V
LTC1628*
*OR TWO LTC1625s
LTC1703
LDO
5V/3A
1.5V/2A
1.3V/10A
3.3V/5A
2.5V/0.5A
1703 F14
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LTC1703
1703fa
efficiency, since a significant fraction of the total power is
drawn from the 3.3V and 5V rails in a typical system. The
correct way to calculate system efficiency is to calculate
the power lost in each stage of the converter, and divide
the total output power from all outputs by the sum of the
output power plus the power lost:
Efficiency
TotalOutputPower
TotalOutputPower TotalPowerLost
=
+
()
100%
In our example 2-step system, the total output power is:
Total output power =
15W + 16.5W + 1.25W + 3W + 13W = 48.75W
corresponding to 5V, 3.3V, 2.5V, 1.5V and 1.3V output
voltages.
Assuming the LTC1703 provides 90% efficiency at each
output, the additional load on the 5V and 3.3V supplies is:
1.3V: 13W/90% = 14.4W/3.3V = 4.4A from 3.3V
1.5V: 3W/90% = 3.3W/5V = 0.67A from 5V
2.5V: 1.25W/75% = 1.66W/3.3V = 0.5A from 3.3V
If the 5V and 3.3V supplies are each 94% efficient, the
power lost in each supply is:
1.3V: 14.4W – 13W = 1.4W
1.5V: 3.3W – 3W = 0.3W
2.5V: 1.66W – 1.25W = 0.4W
3.3V: 16.5W + 3.3V (4.4A + 0.5A) = 32.67W load
(32.67W/94%) – 32.67W = 2.09W lost
5V: 15W + 5V (0.67A) = 18.4W load
(18.4W/94%) – 18.4W = 1.17W lost
Total loss = 5.36W
Total system efficiency =
48.75W/(48.75W + 5.36W) = 90.1%
Maximizing High Load Current Efficiency
Efficiency at high load currents (when the LTC1703 is
operating in continuous mode) is primarily controlled by
the resistance of the components in the power path
(QT, QB, L
EXT
) and power lost in the gate drive circuits due
to MOSFET gate charge. Maximizing efficiency in this
region of operation is as simple as minimizing these
terms.
The behavior of the load over time affects the efficiency
strategy. Parasitic resistances in the MOSFETs and the
inductor set the maximum output current the circuit can
supply without burning up. A typical efficiency curve
(Figure 15) shows that peak efficiency occurs near 30% of
this maximum current. If the load current will vary around
the efficiency peak and will spend relatively little time at the
maximum load, choosing components so that the average
load is at the efficiency peak is a good idea. This puts the
maximum load well beyond the efficiency peak, but usu-
ally gives the greatest system efficiency over time, which
translates to the longest run time in a battery-powered
system. If the load is expected to be relatively constant at
the maximum level, the components should be chosen so
that this load lands at the peak efficiency point, well below
the maximum possible output of the converter.
LOAD CURRENT (A)
0
70
EFFICIENCY (%)
80
90
100
510
1703 G01
15
V
IN
= 5V
V
OUT
= 3.3V
V
OUT
= 2.5V
V
OUT
= 1.6V
Figure 15. Typical LTC1703 Efficiency Curves
Maximizing Low Load Current Efficiency
Low load current efficiency depends strongly on proper
operation in discontinuous and Burst Mode operations. In
an ideally optimized system, discontinuous mode reduces
conduction losses but not switching losses, since each
power MOSFET still switches on and off once per cycle. In
a typical system, there is additional loss in discontinuous
mode due to a small amount of residual current left in the
inductor when QB turns off. This current gets dissipated
across the body diode of either QT or QB. Some LTC1703
systems lose as much to body diode conduction as they
save in MOSFET conduction. The real efficiency benefit of
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LTC1703IG#TRPBF

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Buy LTC1703IG#TRPBF
Manufacturer:
Analog Devices / Linear Technology
Description:
Switching Voltage Regulators 2x 550kHz Sync 2-PhSw Reg Cntr w/ 5-B VI
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