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LT8310IFE#TRPBF

P1-P3P4-P6P7-P9P10-P12P13-P15P16-P18P19-P21P22-P24P25-P27P28-P30P31-P33P34-P36
LT8310
25
8310f
For more information www.linear.com/LT8310
applicaTions inForMaTion
switching frequencies, which leads to smaller magnetics.
The required switching current, I
GATE
, can be calculated
using Equation 32, see the Thermal Considerations section
for further details.
I
GATE
= Q
G
• f
SW
[32]
The power dissipated in the primary MOSFET in a forward
converter is described by Equation 33. The first term
represents the conduction loss in the device, and the
second term represents the switching loss. C
RSS
is the
reverse-transfer capacitance, which is usually specified
in the MOSFET characteristics. For maximum efficiency,
R
DS(ON)
and C
RSS
should be minimized.
P
SW
=I
2
L(MAX)
•R
DS(ON)
•D
MAX
+2• V
2
IN
• C
RSS
• f
SW
•
I
L(MAX)
1A
[33]
From the known power dissipated in the main MOSFET,
its junction temperature can be obtained using Equation
34. T
J
must not exceed the MOSFET maximum junction
temperature rating. It is recommended to measure the
MOSFET temperature in steady state to ensure that absolute
maximum ratings are not exceeded.
T
J
= T
A
+ P
SW
• θ
JA
= T
A
+ P
SW
• (θ
JC
+ θ
CA
) [34]
Input Capacitor Selection
The input capacitor supplies the transient input current
through to the transformer and main switch, so it must be
sized according to transient current requirements. Forward
converters experience discontinuous input currents on par
with the load current divided by the transformer turns ratio.
The switching frequency, output current, and tolerable input
voltage ripple are key inputs to estimating the capacitor
value required to limit input voltage ripple to a specified
level. An X7R type ceramic capacitor is usually the best
choice since it has the least variation with temperature and
DC bias. Low ESR and ESL at the switching frequency are
necessary to avoid excess spiking of the input voltage.
To achieve RMS input ripple of V
IN(RIPPLE)
, the input
capacitor for a forward converter can be estimated using
Equation 35. For example, 15µF is
an appropriate
selec-
tion for 100
mV RMS ripple on a 350kHz converter with
2A maximum load current and a transformer turns ratio
of N
P
/N
S
= 2.
C
IN
=
0.5 • I
L(MAX)
f
SW
• V
IN(RIPPLE)
• N
P
/N
S
( )
[35]
Table 3 provides some recommended ceramic capacitor
vendors.
Table 3.Recommended Ceramic Capacitor Manufacturers
MANUFACTURER WEB ADDRESS
Kemet www.kemet.com
Murata www.murata.com
Taiyo Yuden www.t-yuden.com
TDK www.tdk.com
Inductor Selection
The inductor used with the LT8310 should have a saturation
current rating appropriate to the maximum load current, and
thus appropriate to the switch current rating and R
SENSE
resistor. For applications with no output voltage feedback,
choose an inductor value that keeps ripple current low in
support of the minimum load current target, I
L(MIN)
. If
the contribution of the output inductor equals that of the
reflected transformer magnetizing inductance (Lµ), a first
cut for the inductor value based on operating frequency,
output voltage, and minimum duty cycle is:
L1=
V
OUT
• 1−D
MIN
( )
f
SW
• I
L(MIN)
[36]
Once both the transformer and inductor are chosen, the
minimum load current estimate in Equation 27 should be
re-evaluated, and the component selections modified if
necessary.
LT8310
26
8310f
For more information www.linear.com/LT8310
applicaTions inForMaTion
For applications where current mode control dominates,
choose an inductor value that provides a current mode
ramp on SENSE during the switch on-time of approximately
20mV magnitude based on operating frequency, output
voltage, minimum duty cycle and transformer turns ratio.
The following equation is useful to estimate the inductor
value for continuous conduction mode operation:
L1=
V
OUT
• 1
−
D
MIN
( )
f
SW
•
R
SENSE
N
P
/N
S
( )
• 20mV
[37]
Table 4 provides some recommended inductor vendors.
Table 4. Recommended Inductor Manufacturers
MANUFACTURER WEB ADDRESS
Champs Technologies www.champs-tech.com
Coilcraft www.coilcraft.com
Cooper-Coiltronics www.cooperet.com
Vishay www.vishay.com
Würth-Midcom www.we-online.com
Secondary-Side Switch Selection
A nonsynchronous application, with or without output
voltage feedback, requires only Schottky diode switches
in the secondary. The forward diode conducts the full
(increasing) inductor current when the primary switch
is closed, and the reflected magnetization current (much
smaller) after resonant reset completes. The catch diode
conducts the full (decreasing) inductor current when the
main switch turns off, which is reduced by the magnetiza
-
tion current after resonant reset completes (see Figure 3).
Three-pin dual-packaged diodes may be used to save board
space because the diodes share a node, but the switches
see different reverse voltages, which may favor different
parts in higher current applications. The forward diode
must withstand in reverse the full primary switch node
resonance voltage divided by the primary-to-secondary
turns ratio (N
P
/N
S
); see Equation 30 for an estimate of
the resonant maximum. The catch diode must withstand
the maximum input voltage divided by the turns ratio
in reverse. However in step-up applications, the catch
node may ring, which would require a higher rating for
the switch, or a snubber to limit the peak voltage. When
choosing
diode breakdown ratings consider the likelihood
of abnormal operating conditions. For example: incomplete
resonant reset increasing the switch node voltage and
reverse stress on the forward diode, or sub minimum load
current resulting in increased output voltage and reverse
stress on the catch diode.
As in any converter, the voltage drop across the switches
reduces efficiency, which is reason enough to use low
threshold Schottky diodes with low series resistance.
In duty mode control dominated applications, the actual
output voltage is reduced from the target voltage by the
diode drop. The nominal forward voltage drop at a fixed
load can be planned into the target voltage if desired (see
the section, Programming the Duty Loop Output Voltage
Target. Both the forward and catch diodes must be rated
for the maximum inductor current, have suitable power
dissipation ratings, and be fast enough relative to the
switching frequency to achieve crisp turn-on and turn-
off edges.
Table 5 provides some recommended diode vendors.
Table 5. Recommended Diode Manufacturers.
MANUFACTURER WEB ADDRESS
Central Semiconductor www.centralsemi.com
Diodes, Inc. www.diodes.com
ON Semiconductor www.onsemi.com
Vishay www.vishay.com
Synchronous applications with MOSFET switches in the
secondary have the same stresses and requirements as
diodes, but the advantage of smaller forward voltage drops.
The LT8310 provides the non-overlapping SOUT signal
that is the inverse of the GATE drive for synchronizing
switch drivers such as the LT8311 or LTC3900 to avoid
cross-conduction, see their data sheets for details.
Synchronous switches will experience body diode con
-
duction at
start-up, shutdown, and during small delays
each
switching period. Consider body diode current and
reverse recovery time when selecting MOSFET switches.
LT8310
27
8310f
For more information www.linear.com/LT8310
applicaTions inForMaTion
Output Capacitor Selection
The inductor in the output stage of a forward converter
ensures continuous load current, hence for constant or
slowly varying loads, the output capacitance has a relatively
easy task of filtering inductor ripple current. Fast load
steps withdraw or deposit capacitor charge that changes
the output voltage until inductor current reacts to restore
it and meet the new load demand.
In current mode control applications, tight coupling
between the voltage and current feedback loops and the
compensation zero at the V
C
pin make for excellent load
regulation. The recommended output capacitors for these
circuits are a 220µF electrolytic in parallel with a small
X7R type ceramic capacitor with low equivalent series
resistance (ESR).
In duty mode control applications, no load voltage feed
-
back is present, so the peak transient output excursion
(ΔV
OUT(PK)
) goes as the product of the L-C filter output
impedance (√L1/C
L
), and the magnitude of the load cur-
rent step (Δ
I
L(MAX)
). Assuming L1 is fixed by other con-
siderations, maximize
the load capacitance to minimize
the transient peak, as shown in Equation 38. The ESR
specification of the capacitor should be chosen to satisfy
Equation 39, to minimize
its effect.
Arrange multiple X7R type ceramic capacitors in paral-
lel to
achieve very low ESR and the desired amount of
capacitance
with good temperature and bias stability.
Substituting a high valued electrolytic with high ESR in
parallel with a small X7R capacitor does not provide the
same performance, and should be avoided.
C
L
≥
ΔI
L(MAX)
ΔV
OUT(PK)
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟
2
• L
1
[38]
ESR
2
L1
C
L
[39]
In steady-state, output voltage ripple arises from induc-
tor ripple current that charges and discharges the output
capacitor,
and from the voltage drop across its ESR.
Equation 40 provides an estimate of the output ripple in
relation to the nominal output voltage.
V
OUT(RIPPLE)
≈
1
L1• C
L
• f
2
SW
+
ESR
L1• f
SW
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟
• V
OUT(NOM)
[40]
Programming the Output Voltage in Direct-Wired
Feedback Applications
For nonisolated applications, direct-wired feedback from
the load to the FBX pin configures the LT8310 as a tra
-
ditional peak
current mode controlled forward converter.
The
FBX pin features dual references (1.6V and –0.8V)
that support DC/DC conversion or DC/DC inversion auto
-
matically. Proper selection of the transformer turns ratio
also makes large conversion/inversion ratios (step-down
or step-up) possible without relying on extremely low or
high duty cycles, which improves efficiency. In wired ap
-
plications, the
output voltage (V
OUT
) is set by a resistor
divider, as shown in Figure 8.
Figure 8. Wired Feedback for Nonisolated
Supply Applications
LT8310
V
INTVCC
V
OUT
8310 F08
V
C
R
Z
C
C
R6
R5
DFILT
INTV
CC
RDVIN
FBX
GND
R
SET
C
DFILT
C0
O P T.
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LT8310IFE#TRPBF

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Description:
Switching Voltage Regulators 100Vin For Conv Cntr
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