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LT3431IFE#TRPBF

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LT3431
sn3431 3431fs
APPLICATIO S I FOR ATIO
WUUU
time involved in both the current comparator and turnoff
of the output switch. These result in a minimum on time
t
ON(MIN)
. When combined with the large ratio of V
IN
to
(V
F
+ I • R), the diode forward voltage plus inductor I • R
voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:
ft
VIR
V
ON
F
IN
•
•
≤
+
where:
f = switching frequency
t
ON
= switch minimum on time
V
F
= diode forward voltage
V
IN
= Input voltage
I • R = inductor I • R voltage drop
If this condition is not observed, the current will not be
limited at I
PK
, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT3431 clock frequency
of 500KHz, a V
IN
of 12V and a (V
F
+ I • R) of say 0.6V, the
maximum t
ON
to maintain control would be approximately
100ns, an unacceptably short time.
The solution to this dilemma is to slow down the oscilla-
tor when the FB pin voltage is abnormally low thereby
indicating some sort of short-circuit condition. Oscillator
fre
quency is unaffected until FB voltage drops to about 2/3
of its normal value. Below this point the oscillator fre-
quency decreases roughly linearly down to a limit of about
100kHz. This lower oscillator frequency during short-
circuit conditions can then maintain control with the
effective minimum on time. Even with frequency foldback,
however, the LT3431 will not survive a permanent output
short at the absolute maximum voltage rating of V
IN
= 60V;
this is defined solely by internal semiconductor junction
breakdown effects.
For the maximum input voltage allowed during an output
short to ground, the previous equation defining minimum
on-time can be used. Assuming V
F
(D1 catch diode) =
0.52V at 2.5A (short-circuit current is folded back to
typical switch current limit • 0.5), I (inductor) • DCR = 2.5A
• 0.027 = 0.068V (L = UP2B-100), typical f = 100kHz
(folded back) and typical minimum on-time = 275ns, the
maximum allowable input voltage during an output short
to ground is typically:
V
IN
= (0.52V + 0.068V)/(100kHz • 275ns)
V
IN(MAX)
= 21V
Increasing the DCR of the inductor will increase the
maximum V
IN
allowed during an output short to ground
but will also drop overall efficiency during normal opera-
tion.
It is recommended that for [V
IN
/(V
OUT
+ V
F
)] ratios > 4, a
soft-start circuit should be used to control the output
capacitor charge rate during start-up or during recovery
from an output short circuit, thereby adding additional
control over peak inductor current. See Buck Converter
with Adjustable Soft-Start later in this data sheet.
OUTPUT CAPACITOR
The LT3431 will operate with either ceramic or tantalum
output capacitors. The output capacitor is normally cho-
sen by its effective series resistance (ESR), because this
is what determines output ripple voltage. The ESR range
for typical LT3431 applications using a tantalum output
capacitor is 0.05Ω to 0.2Ω. A typical output capacitor is an
AVX type TPS, 100µF at 10V, with a guaranteed ESR less
than 0.1Ω. This is a “D” size surface mount solid tantalum
capacitor. TPS capacitors are specially constructed and
tested for low ESR, so they give the lowest ESR for a given
volume. The value in microfarads is not particularly criti-
cal, and values from 22µF to greater than 500µF work well,
but you cannot cheat mother nature on ESR. If you find a
tiny 22µF solid tantalum capacitor, it will have high ESR,
and output ripple voltage will be terrible. Table 3 shows
some typical solid tantalum surface mount capacitors.
Table 3. Surface Mount Solid Tantalum Capacitor ESR
and Ripple Current
ESR (Max.,
Ω
) Ripple Current (A)
E Case Size
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
D Case Size
AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
C Case Size
AVX TPS 0.2 (typ) 0.5 (typ)
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LT3431
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APPLICATIO S I FOR ATIO
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Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true, and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the
output
capacitor. Solid
tantalum capacitors fail during very high
turn-on
surges,
which do not occur at the output of regulators. High
discharge
surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
Unlike the input capacitor, RMS ripple current in the
output capacitor is normally low enough that ripple cur-
rent rating is not an issue. The current waveform is
triangular with a typical value of 250mA
RMS
. The formula
to calculate this is:
Output capacitor ripple current (RMS):
I
VVV
LfV
RIPPLE RMS
OUT IN OUT
IN
()
=
()
−
()
()()( )
029.
Ceramic Capacitors
Ceramic capacitors are generally chosen for their good
high frequency operation, small size and very low ESR
(effective series resistance). Their low ESR reduces out-
put ripple voltage but also removes a useful zero in the
loop frequency response, common to tantalum capaci-
tors. To compensate for this, a resistor R
C
can be placed
in series with the V
C
compensation capacitor C
C
. Care
must be taken however, since this resistor sets the high
frequency gain of the error amplifier, including the gain at
the switching frequency. If the gain of the error amplifier
is high enough at the switching frequency, output ripple
voltage (although smaller for a ceramic output capacitor)
may still affect the proper operation of the regulator. A
filter capacitor C
F
in parallel with the R
C
/C
C
network is
suggested to control possible ripple at the V
C
pin. An “All
Ceramic” solution is possible for the LT3431 by choosing
the correct compensation components for the given
application.
Example: For V
IN
= 8V to 20V, V
OUT
= 5V at 2A, the LT3431
can be stabilized, provide good transient response and
maintain very low output ripple voltage using the follow-
ing component values: (refer to the first page of this data
sheet for component references) C3 = 2.2µF, R
C
= 1.5k,
C
C
= 15nF, C
F
= 220pF and C1 = 47µF. See Application
Note 19 for further detail on techniques for proper loop
compensation.
INPUT CAPACITOR
Step-down regulators draw current from the input supply
in pulses. The rise and fall times of these pulses are very
fast. The input capacitor is required to reduce the voltage
ripple this causes at the input of LT3431 and force the
switching current into a tight local loop, thereby minimiz-
ing EMI. The RMS ripple current can be calculated from:
IIVVVV
RIPPLE RMS
OUT OUT IN OUT IN
()
=
()
–/
2
Ceramic capacitors are ideal for input bypassing. At 500kHz
switching frequency, the energy storage requirement of
the input capacitor suggests that values in the range of
2.2µF to 10µF are suitable for most applications. If opera-
tion is required close to the minimum input required by the
output of the LT3431, a larger value may be required. This
is to prevent excessive ripple causing dips below the
minimum operating voltage resulting in erratic operation.
Depending on how the LT3431 circuit is powered up you
may need to check for input voltage transients.
The input voltage transients may be caused by input
voltage steps or by connecting the LT3431 converter to an
already powered up source such as a wall adapter. The
sudden application of input voltage will cause a large
surge of current in the input leads that will store energy in
the parasitic inductance of the leads. This energy will
cause the input voltage to swing above the DC level of input
power source and it may exceed the maximum voltage
rating of input capacitor and LT3431.
The easiest way to suppress input voltage transients is to
add a small aluminum electrolytic capacitor in parallel with
the low ESR input capacitor. The selected capacitor needs
to have the right amount of ESR in order to critically
dampen the resonant circuit formed by the input lead
inductance and the input capacitor. The typical values of
ESR will fall in the range of 0.5Ω to 2Ω and capacitance will
fall in the range of 5µF to 50µF.
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LT3431
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APPLICATIO S I FOR ATIO
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If tantalum capacitors are used, values in the 22µF to
470µF range are generally needed to minimize ESR and
meet ripple current and surge ratings. Care should be
taken to ensure the ripple and surge ratings are not
exceeded. The AVX TPS and Kemet T495 series are surge
rated. AVX recommends derating capacitor operating
voltage by 2:1 for high surge applications.
CATCH DIODE
Highest efficiency operation requires the use of a Schottky
type diode. DC switching losses are minimized due to its
low forward voltage drop, and AC behavior is benign due
to its lack of a significant reverse recovery time. Schottky
diodes are generally available with reverse voltage ratings
of up to 60V and even 100V, and are price competitive with
other types.
The use of so-called “ultrafast” recovery diodes is gener-
ally not recommended. When operating in continuous
mode, the reverse recovery time exhibited by “ultrafast”
diodes will result in a slingshot type effect. The power
internal switch will ramp up V
IN
current into the diode in an
attempt to get it to recover. Then, when the diode has
finally turned off, some tens of nanoseconds later, the V
SW
node voltage ramps up at an extremely high dV/dt, per-
haps 5 to even 10V/ns! With real world lead inductances,
the V
SW
node can easily overshoot the V
IN
rail. This can
result in poor RFI behavior and if the overshoot is severe
enough, damage the IC itself.
The suggested catch diode (D1) is an International Recti-
fier 30BQ060 Schottky. It is rated at 3A average forward
current and 60V reverse voltage. Typical forward voltage
is 0.52V at 3A. The diode conducts current only during
switch off time. Peak reverse voltage is equal to regulator
input voltage. Average forward current in normal opera-
tion can be calculated from:
I
IVV
V
D AVG
OUT IN OUT
IN
()
–
=
()
This formula will not yield values higher than 3A with
maximum load current of 3A.
BOOST␣ PIN␣
For most applications, the boost components are a 0.22µF
capacitor and a MMSD914TI diode. The anode is typically
connected to the regulated output voltage to generate a
voltage approximately V
OUT
above V
IN
to drive the output
stage. However, the output stage discharges the boost
capacitor during the on time of the switch. The output
driver requires at least 3V of headroom throughout this
period to keep the switch fully saturated. If the output
voltage is less than 3.3V, it is recommended that an
alternate boost supply is used. The boost diode can be
connected to the input, although, care must be taken to
prevent the 2× V
IN
boost voltage from exceeding the
BOOST pin absolute maximum rating. The additional
voltage across the switch driver also increases power
loss, reducing efficiency. If available, an independent
supply can be used with a local bypass capacitor.
A 0.22µF boost capacitor is recommended for most appli-
cations. Almost any type of film or ceramic capacitor is
suitable, but the ESR should be <1Ω to ensure it can be
fully recharged during the off time of the switch. The
capacitor value is derived from worst-case conditions of
1840ns on time, 75mA boost current and 0.7V discharge
ripple. The boost capacitor value could be reduced under
less demanding conditions, but this will not improve
circuit operation or efficiency. Under low input voltage and
low load conditions, a higher value capacitor will reduce
discharge ripple and improve start-up operation.
SHUTDOWN FUNCTION AND UNDERVOLTAGE
LOCKOUT
Figure 4 shows how to add undervoltage lockout (UVLO)
to the LT3431. Typically, UVLO is used in situations where
the input supply is
current limited
, or has a relatively high
source resistance. A switching regulator draws constant
power from the source, so source current increases as
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.
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LT3431IFE#TRPBF

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Switching Voltage Regulators Hi V, 3A, 500kHz Buck Sw Reg
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