NCP81241MNTXG Datasheet NCP81241MNTXG | P19-P21

NCP81241

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Remote Sense Amplifier

A high performance high input impedance true

differential amplifier is provided to accurately sense the

output voltage of the regulator. The VSP and VSN inputs

should be connected to the regulator’s output voltage sense

points. The remote sense amplifier takes the difference of

the output voltage with the DAC voltage and adds the droop

voltage to

DIFOUT

VSP

* V

VSN

)

1.3 V * V

DAC

)

CSCOMP

* V

CSREF

This signal then goes through a standard error

compensation network and into the inverting input of the

error amplifier. The non−inverting input of the error

amplifier is connected to the same 1.3 V reference used for

the differential sense amplifier output bias.

Remote Sense Amplifier

The differential current−sense circuit diagram is shown in

the figure below. An internally−used voltage signal Vcs,

representing the inductor current level, is the voltage

difference between CSREF and CSCOMP. The output side

of the inductor is used to create a low impedance virtual

ground. The current−sense amplifier actively filters and

gains up the voltage applied across the inductor to recover

the voltage drop across the inductor’s DC resistance(DCR).

RCS_NTC is placed close to the inductor to sense the

temperature. This allows the filter time constant and gain to

be a function of the Rth_NTC resistor and compensate for

the change in the DCR with temperature.

The DC gain in the current sensing loop is

GCS = VCS/VDCR = (VCSREF−VSCOMP) / (Iout *

DCR) = RCS/RCS3

Where

RCS=RCS2+((RCS1*RCS_NTC)/(RCS1+RCS_NTC))

High Performance Voltage Error Amplifier

A high performance error amplifier is provided for high

bandwidth transient performance. A standard type 3

compensation circuit is normally used to compensate the

system.

Current Sense Amplifier

The outut current signal is floating with respect to CSREF.

The current signal is the difference between CSCOMP and

CSREF. The output side of the inductor is used to create a

low impedance virtual ground. The amplifier actively filters

and gains up the voltage applied across the inductor to

recover the voltage drop across the inductor series resistance

(DCR). Rth is placed near the inductor to sense the

temperature of the inductor. This allows the filter time

constant and gain to be a function of the Rth NTC resistor

and compensate for the change in the DCR with

temperature.

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The DC gain equation for the current sensing:

CSCOMP−CSREF

Rcs2 )

Rcs1@Rth

Rcs1)Rth

Rph

Iout

Total

@ DCR

Set the gain by adjusting the value of the Rph resistor. The

DC gain should be set to the output voltage droop. If the

voltage from CSCOMP to CSREF is less than 100 mV at

ICCMAX then it is recommend increasing the gain of the

CSCOMP amp. This is required to provide a good current

signal to offset voltage ratio for the ILIMIT pin. When no

droop is needed, the gain of the amplifier should be set to

provide ~100 mV across the current limit programming

resistor at full load. The values of Rcs1 and Rcs2 are set

based on the 100 k NTC and the temperature effect of the

inductor and should not need to be changed. The NTC

should be placed close to the inductor.

The pole frequency in the CSCOMP filter should be set

equal to the zero from the output inductor. This allows the

circuit to recover the inductor DCR voltage drop current

signal. Ccs1 and Ccs2 are in parallel to allow for fine tuning

of the time constant using commonly available values. It is

best to fine tune this filter during transient testing.

DCR@25C

2 @ PI @ L

Phase

Programming Current Limit

The current limit thresholds are programmed with a

resistor between the ILIMIT and CSCOMP pins. The

ILIMIT pin mirrors the voltage at the CSREF pin and

mirrors the sink current internally to IOUT (reduced by the

IOUT Current Gain) and the current limit comparators. The

100% current limit trips if the ILIMIT sink current exceeds

10 mA for 50 ms. The 150% current limit trips with minimal

delay if the ILIMIT sink current exceeds 15 mA. Set the

value of the current limit resistor based on the

CSCOMP−CSREF voltage as shown below. To recover

from an OCP fault the EN pin must be cycled low.

LIMIT

2 @

Rcs2

)

Rcs1@Rth

Rcs1)Rth

Rph

Iout

LIMIT

@ DCR

10 m

LIMIT

2 @ V

CSCOMP−CSREF@ILIMIT

10 m

Programming IOUT

The IOUT pin sources a current in proportion to the

ILIMIT sink current. The voltage on the IOUT pin is

monitored by the internal A/D converter and should be

scaled with an external resistor to ground such that a load

equal to ICCMAX generates a 2 V signal on IOUT. A

pull−up resistor from 5 V VCC can be used to offset the

IOUT signal positive if needed.

IOUT

2.0 V

LIMIT

10 @

Rcs2)

Rcs1@Rth

Rcs1)Rth

Rph

Iout

ICC_MAX

@ DCR

@ 2

Programming ICC_MAX

The SVID interface provides the platform ICC_MAX

value at register 21h. A resistor to ground on the IMAX pin

programs these registers at the time the part is enabled.

10 mA is sourced from these pins to generate a voltage on the

program resistor. The value of the register is 1 A per LSB and

is set by the equation below. The resistor value should be no

less than 10 k.

ICC_MAX

21h

R @ 10 mA @ 64 A

Programming TSENSE

A temperature sense inputs are provided. A precision

current is sourced out the output of the TSENSE pin to

generate a voltage on the temperature sense network. The

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voltage on the temperature sense input is sampled by the

internal A/D converter. A 100 k NTC similar to the VISHAY

ERT−J1VS104JA should be used. Rcomp1 is mainly used

for noise. See the specification table for the thermal sensing

voltage thresholds and source current.

AGND AGND

Cfilter

10 nF

TSENSE

Rcomp1

0.0

Rcomp2

8.2 K

RNTC

100 K

Precision Oscillator

Switching frequency is programmed by a resistor ROSC

to ground at the ROSC pin. The typical frequency range is

from 500 KHz to 1.2 MHz. The FREQ pin provides

approximately 2 V out and the source current is mirrored

into the internal ramp generator. The switching frequency

can be found in figure below with a given ROSC. The

frequency shown in the figure is under condition of 10 A

output current at VID = 1 V.

Figure 6. Switching Frequency vs. R

FREQ

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