AD8307 Data Sheet
Rev. E | Page 12 of 24
01082-026
IN
V
LIM
I
OUT
A/0
g
m
A/0
g
m
A/0
g
m
A/0
g
m
g
m
AV
IN
A
2
V
IN
A
3
V
IN
A
4
V
IN
Figure 26. Log Amp Using A/0 Stages and Auxiliary Summing Cells
The chief advantage of this approach is that the slope voltage can
now be decoupled from the knee voltage, E
K
= 2 kT/q, which is
inherently PTAT. By contrast, the simple summation of the cell
outputs results in a very high temperature coefficient of the
slope voltage given in Equation 6. To do this, the detector stages
are biased with currents (not shown), which are rendered stable
with temperature. These are derived either from the supply voltage
(as in the AD606 and AD608) or from an internal band gap
reference (as in the AD640 and AD8307). This topology affords
complete control over the magnitude and temperature behavior
of the logarithmic slope, decoupling it completely from E
K
.
A further step is needed to achieve the demodulation response,
required when the log amp converts an alternating input into a
quasi-dc baseband output. This is achieved by altering the g
m
cells used for summation purposes to also implement the rectifica-
tion function. Early discrete log amps based on the progressive
compression technique used half-wave rectifiers. This made
postdetection filtering difficult. The AD640 was the first
commercial monolithic log amp to use a full-wave rectifier, a
practice followed in all subsequent Analog Devices types.
These detectors can be modeled as essentially linear g
m
cells, but
produce an output current independent of the sign of the voltage
applied to the input of each cell; that is, they implement the
absolute value function. Because the output from the later A/0
stages closely approximates an amplitude symmetric square
wave for even moderate input levels (most stages of the amplifier
chain operate in a limiting mode), the current output from
each detector is almost constant over each period of the input.
Somewhat earlier detector stages produce a waveform having
only very brief dropouts, whereas the detectors nearest the
input produce a low level, almost sinusoidal waveform at twice
the input frequency. These aspects of the detector system result
in a signal that is easily filtered, resulting in low residual ripple
on the output.
INTERCEPT CALIBRATION
All monolithic log amps from Analog Devices include accurate
means to position the intercept voltage, V
X
(or equivalent power for
a demodulating log amp). Using the scheme shown in Figure 26,
the basic value of the intercept level departs considerably from
that predicted by the simpler analyses given earlier. However,
the intrinsic intercept voltage is still proportional to E
K
, which is
PTAT (see Equation 5). Recalling that the addition of an offset to
the output produces an effect that is indistinguishable from a
change in the position of the intercept, it is possible to cancel
the left-right motion of V
X
resulting from the temperature
variation of E
K
. Do this by adding an offset with the required
temperature behavior.
The precise temperature shaping of the intercept positioning offset
results in a log amp having stable scaling parameters, making it a
true measurement device, for example, as a calibrated received
signal strength indicator (RSSI). In this application, the user is
more interested in the value of the output for an input waveform
that is invariably sinusoidal. Although the input level can alterna-
tively be stated as an equivalent power, in dBm, be sure to work
carefully. It is essential to know the load impedance in which
this power is presumed to be measured.
In radio frequency (RF) practice, it is generally safe to assume a
reference impedance of 50 in which 0 dBm (1 mW) corresponds
to a sinusoidal amplitude of 316.2 mV (223.6 mV rms). The
intercept can likewise be specified in dBm. For the AD8307, it is
positioned at −84 dBm, corresponding to a sine amplitude of 20 µV.
It is important to bear in mind that log amps do not respond to
power, but to the voltage applied to their input.
The AD8307 presents a nominal input impedance much higher
than 50 (typically 1.1 k low frequencies). A simple input
matching network can considerably improve the sensitivity of
this type of log amp. This increases the voltage applied to the
input and thus alters the intercept. For a 50 match, the voltage
gain is 4.8 and the entire dynamic range moves down by 13.6 dB
(see Figure 35). Note that the effective intercept is a function of
waveform. For example, a square wave input reads 6 dB higher
than a sine wave of the same amplitude and a Gaussian noise
input 0.5 dB higher than a sine wave of the same rms value.
OFFSET CONTROL
In a monolithic log amp, direct coupling between the stages is
used for several reasons. First, this avoids the use of coupling
capacitors, which typically have a chip area equal to that of a
basic gain cell, thus considerably increasing die size. Second, the
capacitor values predetermine the lowest frequency at which the
log amp can operate; for moderate values, this can be as high as
30 MHz, limiting the application range. Third, the parasitic
(backplate) capacitance lowers the bandwidth of the cell, further
limiting the applications.
However, the very high dc gain of a direct-coupled amplifier
raises a practical issue. An offset voltage in the early stages of
the chain is indistinguishable from a real signal. For example, if
it were as high as 400 V, it would be 18 dB larger than the
smallest ac signal (50 V), potentially reducing the dynamic
range by this amount. This problem is averted by using a global
feedback path from the last stage to the first, which corrects this
offset in a similar fashion to the dc negative feedback applied
around an op amp. The high frequency components of the
signal must be removed to prevent a reduction of the HF gain in
the forward path.
